Circuits and Systems, 2013, 4, 58-66
http://dx.doi.org/10.4236/cs.2013.41010 Published Online January 2013 (http://www.scirp.org/journal/cs)
A Current Bleeding CMOS Mixer Featuring LO
Amplification Based on Current-Reused Topology*
Wah Ching Lee 1,2, Kim Fung Tsang2, Yi Shen2, Kwok Tai Chui2
1Department of Electronic & Information Engineering, The Hong Kong Polytechnic University, Hong Kong, China
2Department of Electronic Engineering, City University of Hong Kong, Hong Kong, China
Email: enwclee@inet.polyu.edu.hk, ee330015@cityu.edu.hk, yishen@cityu.edu.hk, ktchui3@student.cityu.edu.hk
Received August 30, 2012; revised September 30, 2012; accepted October 8, 2012
ABSTRACT
A double balanced Gilbert-cell class-A amplifier bleeding mixer (DBGC CAAB mixer) is proposed and implemented.
The injection current is utilized to amplify the local oscillator (LO) signal to improve the performance of the transcon-
ductor stage. The DBGC CAAB mixer achieves a conversion gain of 17.5 dB at 14 dBm LO power, and the noise fig-
ure is suppressed from 45 dB to 10.7 dB. It is important to stress that the new configuration will not drain additional
power in contrast to the former current bleeding mixers. This topology dramatically relieves the requirement of the LO
power. The DBGC CAAB mixer is implemented by using 0.18-μm RFCMOS technology and operates at the 2.4 GHz
ISM application with 10 MHz intermediate frequency. The power consumption is 12 mA at 1.5 V supply voltage. The
DBGC CAAB mixer features the highest FOM figure within a wide range of LO power.
Keywords: Mixer; Gilbert-Cell; Current Bleeding; Noise; Conversion Gain; Current-Reuse; Class-A Amplifier
1. Introduction
With the rapid proliferation of modern wireless market,
RF transceivers play a more and more important role in
our daily lives. Mixer is one of the key blocks in the sig-
nal chain performing frequency translation before signals
are further processed in the intermediate frequency cir-
cuits. The noise and the gain performance of the mixer,
in general, determine the performance of the whole sys-
tem at large [1]. In past decades, many research scholars
made deep insights into the noise mechanism [2-5] and
gain enhancement [6-8] in mixers. In order to achieve
good performance including gain, noise, isolation and
linearity (higher-order components), Gilbert-cell [9] is
commonly used as the mixer core topology.
Amongst most factors, the conversion gain (CG) and
the third-order intercept point (IP3) of the mixer are the
key elements determining the mixer performance. Both
CG and IP3 are proportional to the square root of the bias
current (Ib1) of the driver stage [10,11], as shown in Fig-
ure 1(a). In order to improve CG and IP3, a simple
method is to increase the bias current of the driver stage
M1, namely Ib1. However, this improvement is at the
expense of other degradations. For instance, the increase
of Ib1 will also increase the currents, Ib2 and Ib3, in the
switch pair M2 and M3 (referred as M2-M3 thereafter).
As a result, the noise contributions from M2-M3 are also
increased [12-14]. In addition, in order to keep all the
transistors working in saturation region, the load resis-
tance ZL should be decreased to avoid too much voltage
drop across them, especially in modern sub-micro CMOS
technology. The load resistance reduction will in turn
cause the gain compression. Hence it is concluded that
increasing the bias current solely of the driver stage is
not an efficient way to improve the overall performance,
including gain, linearity and noise of the mixer.
In order to retain the benefits of increasing the bias
current but without degrading other performances, a cur-
rent bleeding topology, by applying a current source in
(a) (b)
*The work was supported by Research Grants 9220049 from City Uni-
versity of Hong Kong.
Figure 1. Schematic of (a) Conventional single-balanced
mixer; (b) Single-balanced mixer with current bleeding.
C
opyright © 2013 SciRes. CS
W. C. LEE ET AL. 59
parallel with M2-M3, is used [15]. Analysis shows that
such topology [15], [16] facilitates the quiescent current
in the driver stage to be independent from M2-M3. Fig-
ure 1(b) shows the illustrative topology. The bleeding
current source drives a higher driver stage current
through the higher load resistance of M1 by virtue of the
fact that part of the driver bias current is steered from
M2-M3. An additional advantage of using the bleeding
source is that M2-M3 can operate at a lower gate-source
voltage and thus rendering a compact size. Lower gate-
source voltage helps to improve the conversion effi-
ciency as fewer charges are necessary to turn M2-M3 on
and/or off.
In this paper, a double balanced Gilbert-cell class-A
amplifier bleeding mixer (DBGC CAAB mixer) is de-
veloped. By inserting a class-A amplifier in parallel with
M2-M3, the local oscillator (LO) swing is increased to
drive M2-M3 into a hard switching fashion. An addi-
tional advantage is that the bias current of the class-A
amplifier will enhance the bias of the driver stage of the
mixer, hence improving the overall performance. The
present investigation is an extension of our previous
work [17] with a detailed analysis of noise reduction and
gain boosting. Detailed measurement results are also
provided. The paper is organized as follows: Section 2
describes the motivation of this work; Section 3 de-
scribes the topology, the design and the analysis results,
Section 4 summarizes the performance of the developed
mixer by including a FOM comparison and, Section 5
gives the conclusion.
2. Motivation
In the mixer analysis, the drain current of M2-M3 are
presented as:

2
22DGST
I
KV V

2
33DGST
VV
23
(1)
IK (2)
The voltage of Local Oscillator (LO), vLO, is defined
as:
L
OGS GS
vV V
123
(3)
From KCL:
D
DD
I
II
out2 3
(4)
and, for the differential operation of M2-M3, the output
current is:
D
D
I
II (5)
Based on the physics of the MOS device, Equations (1)
and (2) are supported by the following relationships:
1
2
2
D
nox
od
1
2
I
W
LV
odGS T
VVV
KC
(6)
(7)
In order to characterize the relationship of the output
current and the performance of M2-M3, the output cur-
rent is derived as a function of
O
v:
2
1
out
2D
OLO
I
I
L
Kv v
K
  (8)
This formula is derived based on the condition that
both M2 and M3 are at ON state. When 2
L
Oo
,
M2-M3 acts as a hard switch, and the output current, Iout ,
can be modeled as a sgn function:
d
vV


2
2
1
out
1
11 2,2
sgn ,2
D
D
I
I
I


 
(9)
where,
L
Ood
vV
Figure 2 shows the I-V characteristics of M2-M3
graphically. The transconductance of M2-M3, Gm, is
given by:

 

23
23
2mLO mLO
mLO
mLO mLO
gv gv
Gv
g
vgv
(10)
It will be explained in (13) that increasing vLO will im-
prove the mixer performance; hence the dependence of
Gm on vLO needs to be investigated. Gm is derived as a
function of
L
O
v

:
1
21
1
22
mLO D
D
LO
Gv I (11)
I
v
K

Gv
As shown in Figure 2, mLO
reaches its maximum
value as
O approaches 0. When v
L
O increases, v
Gv
mLO
Noise is an important parameter in mixer design and is
now discussed. In general, there are three main noise
decreases and eventually approaches zero.
Figure 2. I-V characteristic and equivalent transconduc-
tance of M2-M3, where δ = vLO/Vod.
Copyright © 2013 SciRes. CS
W. C. LEE ET AL.
60
sources in conventional mixer circuits, namely the trans-
conductor stage, the switch pair M2-M3 and the load. In
Terrovitis [5] and Darabi [2] models, noise contributed to
the output exactly when both switching transistors are at
ON state—indicated as the switch interval Δ (will be dis-
cussed in Section 3). The noise PSD (power spectrum
density), introduced within Δ is presented as:
23
23
mm
mm
gg
gg



,
816
m
PSD KT GKT

 (12)
where K
Boltzmanns constant
T
Absolute temperature
γ
noise coefficient (= 2/3 for long channel device)
23mm
g
g
Trans-conductance of M2-M3
At the zero-crossing point of vLO, both M2 and M3 are
at ON state, thus resulting in non-zero value of 2m
g
and
3m
g
. From (11), large LO voltage swing will diminish
Gm, rendering either 2m
g
or 3m
g
to vanish and thus
suppressing the noise. Noise PSD can further be con-
verted [5] to the relationship of bias current and noise
contribution:
16
π
L
O
I
KT
v


1mL
gre Z
PSD (13)
where IB is the bias current of M2-M3. It is seen from (13)
that, by increasing vLO, Noise PSD is suppressed effi-
ciently. Furthermore, by reducing IB, PSD will also be
suppressed, thus further verifying the current bleeding
principle.
The other key parameter of a mixer is gain. In order to
enhance the gain, the conversion gain (CG) of the mixer
is also analyzed [5].
CG c (14)
where the multiplier c is given by:

sin π
ππ
LO
LO
T
T




2
c (15)
It is seen from (15) that increasing c, and/or gm1, is an
efficient way to increase CG. The implementation will be
discussed in Section 3.4.
Based on the above discussion, the novelties of the
proposed design are:
A1) Novel bleeding source: In order to improve the
gain, linearity and noise, the bias current of the driver
stage should be increased. However, such a direct in-
crease will cause other related problems, including the
high noise contribution from M2-M3 as well as the ex-
cessive voltage drop on the load resistor. It is important
to note that the current bleeding structure improves only
the bias of the driver stage but without increasing the
bias of M2-M3.
A2) LO amplification: Higher amplitude of VLO helps
to minimize the noise PSD from the M2-M3 by reducing
Δ, and at the same time, maximizing
1mL
In this investigation, a Gilbert-cell mixer based on the
current bleeding technique is explored. Inspired from the
benefit of high gain and low noise (referring to (A1) and
(A2)), and taking advantage of the linearity, a class-A
amplifier (referred as Amp thereafter), is implemented as
the bleeding sources to amplify the LO signals. Hence
the class-A amplifier bleeding source is employed to
replace the traditional simple current source. After per-
forming the LO amplification, the DC bias current of the
class-A amplifier is steered to the driver stage of the
mixer to improve the gain, noise and linearity character-
istics. It will be explained that, comparing to the conven-
tional current bleeding mixer, the CAAB structure can
reuse the bleeding current more efficiently.
2πCGg Z to reach the upper limit.
3. Methodology
3.1. Overall Topology
Figure 3 shows the proposed double-balanced Gilbert-
cell mixer incorporating CAAB, namely DBGC CAAB
mixer. As shown in Figure 1(b), taking advantage of the
symmetry, one half of the circuit is used for analysis. In
conventional mixers, the bias current, Ib1, of the driver
stage M1 is the sum of current of Ib2 (of M2) and Ib3 (of
M3), i.e. Ib1 = Ib2 + Ib3. However, in a DBGC CAAB
mixer, the bias current of M1 is the sum of Ib2, Ib3 and Ia,
i.e. Ib1 = Ib2 + Ib3 + Ia. By devising component values for
Amp, VLO of M2-M3 is increased, thus reducing Δ, and
rendering a noise reduction in M2-M3.
3.2. The Amp Design
The Amp in Figure 3 is implemented as shown in Fig-
ure 4. As described in the preceding section, the bias
current Ia is steered into the driver stage M1 of the mixer.
Thus the provision of a uniform current (Ia) is crucial to
the performance of the mixer since any potential large
Figure 3. Schematic of DBGC CAAB mixer.
Copyright © 2013 SciRes. CS
W. C. LEE ET AL. 61
disturbance of Ia may cause gain ripples and non-linear
degradations in the driver stage M1. In this case, since
0.18-μm fabrication process is used, the channel length
of the transistor is small, rendering Ia more and more
susceptible to the channel length modulation [18]:

21DS
V V
1
2
an oxGSTH
W
ICV
L
 (16)
The large output voltage swing normally appearing on
the drain of the transistor will change the VDS dramati-
cally and periodically. In principle, in order to avoid the
channel length modulation, a cascode topology can be
employed. However, this structure will limit the output
voltage swing. An alternative method is to use a longer
channel length. In this investigation, 0.35-μm gate length
is preferred as the operating frequency is not very high
(2.5 GHz).
It is important to stress that, in order to lower the in-
herent noise in the CAAB mixer, the Amp is realized as
a low noise amplifier. In Figure 4, the components Lg,
Cgs, Csr, Lsr and Ls are used for noise matching [19]. In
order to further analyze the amplifier, the equivalent cir-
cuit is shown in Figure 5. By using a large value of Ls
(for isolating RF signals and LO signals), the impedance
Zm can be considered as open. Csr is a relatively large
value capacitor used for blocking the DC current but will
not influence significantly on the series combination of
Cgs and Csr:
111
gs sr
CCC

comb
Vdd
Lg
Ld
Rb
Cbi
Cout
s
L
Ia
LOout
gs
C
Lsr
Csr
LOin Ia
Figure 4. Topology of current bleeding source—Amp (class-
A amplifier).
2
d
i
gs
s
L
gs
C
m
Vg
g
L
s
R
2
s
i
2
ng
i
2
out
i
Figure 5. Equivalent circuits of the class-A amplifier (Amp)
in Figure 4.
The input impedance Zin is then degenerated to be:

1m
ing srsr
comb gs
g
Z
sL LL
sC C
 
8.5 nH, 0.7 pF, 20 pF, 1.2 nH.LCCL
(17)
At resonance, the imaginary part of Zin vanishes, and
the real part is left for 50 matching [19]. The Amp
operates at a DC bias current of 4.5 mA.
3.3. The DBGC CAAB Mixer and
Implementation
The Amp is then designed and analyzed by SpectreRF in
Cadence. Compromising the power consumption, LO
amplification and stability, the gain of the Amp to be
designed is chosen to be about 14 dB. The component
values in Figure 5 are optimized as:
ggssrsr

Figure 6 shows the final characteristics of Amp. Figure
7 shows the voltage waveforms at the input and the out-
put of the Amp at various LO input power levels. An
investigation of the voltage waveforms reveals that the
amplitude of the LO signal is amplified by a factor of
five (5). The achieved input 1 dB compression point
(P1dB,in) is 3 dBm, thus surpassing the traditional per-
formance that P1dB,in < 0 dBm.
-20-15-10-50510
10
11
12
13
14
Amplifier gain (dB)
LO input power (dBm)
P1dB
Figure 6. The simulated voltage gain of the Amp.
0.0 0.1 0.2 0.3 0.4 0.5
0.0
0.5
1.0
1.5
2.0
A f ter am plifier
Time (Nano-Second)
P
LO
= -8 dBm
P
LO
= -14 dBm
P
LO
= -2 dBm
P
LO
= 4 dBmBefore a mplifier
Voltage (V)
Figure 7. The voltage swing at the input and the output of
Amp at varying input power level.
Copyright © 2013 SciRes. CS
W. C. LEE ET AL.
62
The DBGC CAAB mixer is realized by a 0.18-μm
1-poly 6-metal RFCMOS technology. A microphoto-
graph of the device is shown in Figure 8. The chip occu-
pies an area of 1.2 × 1.3 mm2. Under nominal operation,
the mixer extracts 12 mA from the 1.5 V supply. The
chip was bonded to a FR4-PCB board with gold-wire for
measurement. The operating frequency is 2.4 GHz with
10 MHz IF as the output frequency for testing purpose.
3.4. Noise Analysis
In essence, the LO signal is a sinusoid rather than a
square wave, rendering that the current switching to ap-
proximate a soft-switch with switch interval Δ (see Fig-
ure 9). When v
LO < VX, (VX is the threshold voltage of
M2-M3) both transistors of M2-M3 are “ON”, and the
current division ratio is biased dependently such that the
total bias current of M2-M3 is constant: when M2 is bi-
ased in high level, the current flow through M2 increases,
and the current in M3 decreases. When vLO > VX, the cur-
rent will flow through one of the transistors and the other
counterpart is turned off. At this point, the “ON” transis-
tor acts as the cascode stage of the driver stage.
Figure 8. Microphotograph of the implemented DBGC
CAAB mixer.
o
witch interval
x
x
O
I
o
t
t
t
x
x
1
21
5
1
21
5
oisepulsetrain
Figure 9. The influence of vLO on Δ and the resulting noise
contribution.
The output noise component from M2-M3 is dictated
by the relationship in (10) and (12), in which gm2 and gm3
are time-varying transconductances of M2-M3 under
large LO drive. When vLO > VX, either M2 or M3 is cutoff
and gm vanishes. Consequently, Gm, as well as PS D, also
vanish. When vLO < VX, the non-zero G
m proliferates
M2-M3 to contribute noise to the output.
Let PSD = Ssw,within when vLO < VX and PSD = Ssw,outside
when vLO > VX. For analysis purpose, Ssw,within is normal-
ized to a train of pulses [4] operating at a rate 2fLO, as
shown in Figure 9. The width of Δ is VX/λ, where λ is the
slope of LO waveform at the zero-crossing point, and is
given by:
2π
L
OLO
vT (18)
It was illustrated in Section 3.4 that by devising the
Amp, vLO has been amplified by five (5) times, thus the
switch interval Δ2 (after amplification) has been com-
pressed to one fifth of Δ1 (before amplification) (see
Figure 9).
To further analyze the relationship between the noise
PSD and the slope of LO waveform, λ, at the zero-
crossing point, (12) is modified and derived as follows:
32 B
I
L
O
PSDKT T

LO sw
VSf
(19)
From (19), it is analyzed and concluded that the noise
from M2-M3 is reduced by a factor of five (5). Hence it
is analyzed that:
 

Additional noise improvement comes from the in-
creasing bias of the driver stage due to the current bleed-
ing source (here, it is DC bias of the Amp). Based on the
original 1.5 mA DC bias current, the current bleeding
source feeds additional 4.5 mA DC current to the driver
stage. From [18], 11mBgsth
g
2IVV
, consequently,
gm1 is improved by a factor of 4 . For double-balanced
mixer, the single-side band noise figure, NF [5], is rear-
ranged as:
2
33
1
12
11
22
1
44
2
2
g
g
mm
s
GrGR
rgg
NF ccR



 (20)
where γ1 and γ3 represent the noise coefficient of M1 and
M3 respectively. The poly resistance of the gate is indi-
cated as rg1 and rg3. It is seen from (20) that NF is in-
versely proportional to gm1. Thus it is seen that the cur-
rent bleeding improves gm1 dramatically and cones-
quently suppresses the noise figure efficiently.
To examine the performance improvement of the
DGBC CAAB mixer, an identical conventional Gilbert-
quad mixer, without current bleeding—Amp, but having
Copyright © 2013 SciRes. CS
W. C. LEE ET AL.
CS
63
the same transistor size in both the driver stage and the
switch stage, is also designed with same process. Both
mixers operate at 2.4 GHz input frequency with 10 MHz
IF output. Figure 10 shows the noise comparison at LO
power from 20 dBm to 10 dBm. It is observed that the
conventional mixer presents a noise figure in the range of
13 dB to 46 dB whereas the CAAB mixer features a
much lower noise figure, namely from 9 dB to 11.5 dB.
The inset of Figure 10(b) shows the measured noise fig-
ures versus LO power at 10 MHz intermediate frequency.
The measured NF varies from 12.4 dB to 8.7 dB when
the LO power is varied from 20 dBm to 10 dBm.
LO drive. Figure 11(b) intuitively shows the trend of
Lsw,dB (in logarithm scale) due to soft switching. Table 1
lists the reduction of Lsw,lin due to the LO amplitude am-
plification at four different
L
O cases. In the extreme
case that Δ = 0.45 (when LO voltage is very weak), the
gain improvement can be as high as 3 dB. It is also seen
that when Δ is one fourth of the LO period (TLO), which
is the most probably case, Lsw,lin can be improved from
T
Copyright © 2013 SciRes.
3.5. Conversion Gain Analysis
In the CAAB mixer, the conversion gain is affected by
the switch stage M2-M3 and the driver stage. Recapitu-
lated from (14), a Δdecreases,

sin ππ
L
OLO
approaches to unit. As a result, the CG increases until
reaching the upper limit
TT

2π1mL
g
R. Figure 11 il-
lustrates the switch loss, Lsw, versus Δ. Figure 11(a)
shows Lsw,lin, versus Δ due to the non-ideal square wave
0.92 π1mL
g
R
 to

1
0.9962 πmL
g
R
Conventional mixer
110100
. As a result,
0.88 dB CG is gained.
It is shown in Section 2 that gm1 is improved by a fac-
tor of 4, hence an additional 12 dB gain is obtained. Thus,
it is concluded that the current bleeding structure will
boost gm1, provoking a higher CG.
To examine the gain performance of the DBGC
CAAB mixer, a comparison with the conventional mixer
and other published works is shown in Figure 12. For the
conventional mixer, the maximum gain achieved is 11
dB when LO port is fed by 10 dBm signal. In contrast, a
gain of 17.5 dB is achieved when 14 dBm is fed to the
LO port of the DBGC CAAB mixer. Thus, it is con-
10
50
20
30
40
N
o
i
se
F
ac
t
or
(dB)
Unit: dBm
Plo= - 2
0
Plo= - 1
7
Plo= - 1
4
Plo= - 1 1
Plo=-8
Plo=-5
Plo=-2
Plo=1
Plo=4
Plo=7
Plo=10
110100
10
12
14
16
Plo=-20 dBm Plo=-2 dBm
Plo=-17 dBm Plo=1 dBm
Plo=-14 dBm Plo=4 dBm
Plo=-11 dBm Plo=7 dBm
Plo=-8 dBm Plo=10 dBm
Plo=-5 dBm
CAAB xer
Measured
NF
IF=10 MHz
-15 -10-50510
mi
9
10
11
12
Noise Figure (dB)
LO powe r (dBm)
(a)
(b)
Intermediate frequency (MHz)
Figure 10. Noise figures at different LO input power. (a) Conventional mixer; (b) DBGC CAAB mixer; inset: measured NF at
IF = 10 MHz.
0.0 0.10.2 0.3 0.4 0.5 0.6
0.5
0.6
0.7
0.8
0.9
1.0

/ T
LO
sin(

/T
LO
)/

/T
LO
is reduced by 5 times
0.0 0.1 0.2 0.3 0.4 0.5 0.6
-6.0
-5.5
-5.0
-4.5
-4.0
-3.5
-3.0
-2.5
-2.0
-1.5
-1.0
-0.5
0.0
is reduced by 5 times
sin(

/T
LO
)/

/T
LO
(dB)
0.3 dB

/ T
LO
1.75 dB
0.88 dB
3 dB
(a) (b)
Figure 11. The switch loss reduction due to the switch interval Δ shrinkage: (a) Linear scale; (b) Logarithm scale.
W. C. LEE ET AL.
64
Table 1. The quantized switching loss reduction due to the LO amplification in linear and logarithm scale.

sin ππ
L
OLO
TT sin π
gπ
LO
LO
T
T




20lo sinπ
gπ
LO
LO
T
T



20lo
Linear scale Logarithm scale (dB) Logarithm (dB)
Δ/TLO Without Amp With Amp Without Amp With Amp Switching loss reduction
0.15 0.9634 0.99852 0.3239 0.01286 0.3
0.25 0.90032 0.99589 0.912 0.03574 0.88
0.35 0.81033 0.99196 1.827 0.07012 1.75
0.45 0.69865 0.98673 3.115 0.116 3
-20 -15 -10-50510
-20
-15
-10
-5
0
5
10
15
20
25
Conventional mixer
Simulated (this work)
Measured (this work)
[20] Vidojkovic. V
[21] Hermann, C
[22]
LO input power (dB
Conversion gain (dB)
J. Park
m)
-22 -20 -18 -16 -14 -12 -10-8-6-4
0.6
0.7
0.8
0.9
1.0
1.1
IF peak to peak voltage (V)
RF input power (dB m )
IF Vpp
-40 -30 -20 -10010
Figure 12. The conversion gain comparison between the
DBGC CAAB mixer and other works.
cluded that by incorporating the current bleeding LO
Amp, not only the conversion gain is enhanced, but also
the LO power requirement is relieved by more than 20
dB. The comparison with other published works [20-22]
reveals that the DBGC CAAB mixer has the highest gain
but requiring the lowest LO power requirement.
3.6. Linearity, LO Power Leakage and
Operation Bandwidth
The output voltage swing is examined by terminating an
oscilloscope at the IF (10 MHz) port. The maximum
voltage swing (peak-to-peak voltage) was measured to be
1.1 V when the RF port was fed by 11 dBm input power
(see Figure 13).
Figure 14 shows the measured third order intermodu-
lation (IP3) of the DBGC CAAB mixer at different LO
power levels. When the switch pair is fed with a 5 dBm
LO power, the IIP3 obtained is 3 dBm. When the LO
power is 17 dBm, the mixer still has an IIP3 point of 9
dBm. In the DBGC CAAB mixer, the LO amplitude is
amplified by five (5) times. Hence, attention is drawn to
the LO power leakage. By examining the LO power at
RF and IF port, it is found that the LO to IF port leakage
is smaller than 90 dBm. The LO to RF power leakage is
measured to be less than 60 dBm (see Figure 15). The
good LO-IF isolation achieved is attributed to use of the
double-balanced topology.
Figure 13. IF peak-to-peak voltage versus RF input power.
(The result is measured with the termination of oscilloscope
at IF port).
-120
-100
-80
-60
-40
-20
0
20
Plo=-17
Plo=-14
Plo=-8
Output power (dBm)
RF input power (dBm)
Plo = - 5
Figure 14. IIP3 of the DBGC CAAB mixer at different LO
level.
-20 -15 -10-50510
-120
-105
-90
-75
-60
LO power at other ports (dBm)
LO input power (dBm)
LO power at IF port
LO powe r a t R F po r t
Figure 15. LO power leakage to RF port and IF port.
Copyright © 2013 SciRes. CS
W. C. LEE ET AL. 65
The DBGC CAAB mixer is designed for 2.4 GHz ISM
band application. Figure 16 shows the gain performance
at 2.4 - 2.4835 GHz when the IF frequency is 10 MHz. In
the whole band, the conversion gain of the proposed
mixer varies from 19 dB to 21 dB, featuring good gain
flatness.
4. Performance Summary and Comparison
Table 2 summarizes the performances of the DBGC
CAAB mixer. To evaluate the mixer comprehensively, a
benchmarking figure of merit, FOM, is presented:

Gain23 1020
10
10 log1 mW
LO
NF IIPP
DC
FOMP


01 KHzf

(21)
The FOM takes incorporates important parameters in-
cluding Gain, NF, IIP3, LO power, DC power and oper-
ating frequency into considerations. The FOM values are
listed in Table 2 for comparison. From the comparison, it
is found that the DBGC CAAB mixer has the lowest LO
power requirement (17 dBm) and the highest gain (15.7
dB), while maintaining a relatively lowest noise figure of
9.7 dB.
5. Conclusion
A current bleeding with LO amplification mixer based on
current reuse topology is designed and implemented. The
developed double balanced Gilbert-cell class-A amplifier
bleeding mixer (DBGC CAAB mixer) has the highest
conversion gain at the lowest LO power when compared
to mixers formerly investigated. The DBGC CAAB
mixer is implemented by using 0.18-μm CMOS technol-
ogy and operates at the 2.4 GHz ISM application with 10
MHz intermediate frequency. The power consumption is
12 mA at 1.5 V supply voltage. With the novel LO am-
plification and current reuse technique, the mixer fea-
tures an excellent high gain of 17.5 dB at a very low LO
power feeding of 14 dBm. The noise performance is
also good. The DBGC CAAB mixer features a noise fig-
ure of 10.7 dB, thus rendering the resulting noise to be
suppressed to [8.7 dB, 12.4 dB]. In contrast, in the con-
ventional mixer, the noise figure varies from 13 dB to 46
dB at the same LO feed. It is important to point out that,
compared to the other mixer investigations, the DBGC
CAAB mixer features the highest FOM figure within a
wide range of LO power.
16
18
20
22
24
2.40 2.42 2.44 2.46 2.48 2.50
2.47
2.45
2.43
2.41
f
LO
(GHz)
Conversion gain (dB)
f
RF
(GHz)
Conversion gain at IF frequency=10 MHz
2.39
Figure 16. Conversion gain of the DBGC CAAB mixer ver-
sus frequency when IF frequency = 10 MHz.
Table 2. The performance of the DBGC CAAB mixer w.r.t. other works.
Publication Technology RF VDD PDC P
LO Gain NF P1dB IIP3 FOM
CMOS GHz V mW dBmdB dB dBm dBm dB
Darabi [5] (bleed) 0.13 μm 2 1.2 2.4 - 0.5 13.5 1.5 10.5 -
V. Vidojkovic [20] 0.18 μm 2.4 1.8 8.1 2 15.7 12.9 - 1 46.2
C. Hermann [21] 0.13 μm 2.5 0.6 1.6 1 5.4 14.8 9.2 2.8 44
J. Park [22] (bleed) 0.18 μm 5 - 7 1 16.2 9.8 - 5 46.6
P. J. Sulivan [23] 0.8 μm 1.9 5 133 3 9.7 7.8 - 1 34.6
This work 0.18 μm 2.4 1.5 18
17 15.7 10.7 10 9 48.3
This work 0.18 μm 2.4 1.5 18
14 17.5 10.5 12 8 48.0
This work 0.18 μm 2.4 1.5 18 5 23 9.7 10 3.5 47.6
Copyright © 2013 SciRes. CS
W. C. LEE ET AL.
66
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Copyright © 2013 SciRes. CS